High-frequency power source

ABSTRACT

[SOLUTION] A radio-frequency power source 1 includes two DC-RF converting circuits 4A, 4B and an RF combining circuit 5 for combining the outputs from both DC-RF converting circuits 4A, 4B. The DC-RF converting circuits 4A, 4B amplify radio-frequency voltages va, vb inputted from a radio-frequency signal generating circuit 8, and output radio-frequency voltages vPA, vPB. The RF combining circuit 5 outputs radio-frequency voltage vPX at a ratio corresponding to the phase difference θ between the radio-frequency voltages vPA and vPB. A controlling circuit 9 switches the phase difference θ between θ1 and θ2. As a result, the power PX outputted from the RF combining circuit 5 becomes pulsed radio-frequency power having a high level period and a low level period. Since the switching of the phase difference θ can be performed at high speed, it is possible to output pulsed radio-frequency power with a high switching frequency between the first level and the second level.

TECHNICAL FIELD

The present invention relates to a radio-frequency power source used in a plasma processing system, for example.

BACKGROUND ART

The plasma processing system may be used in the following way. A fluorine-based gas and a workpiece, such as semiconductor wafers or liquid crystal substrates, are placed in the chamber of a plasma processing apparatus. In the chamber a pair of electrodes are provided, and radio-frequency power from the radio-frequency power source is provided to the pair of electrodes for generating electric discharge. This discharge generates a plasma of the gas contained, and thin film forming or etching is performed on the workpiece.

Conventionally, as a radio-frequency power source used in a plasma processing system, the following type is known. A radio-frequency power source outputs radio-frequency power of a predetermined frequency, and the output power is pulse-modulated based on pulse modulation control signals having a low frequency than that of the output power. In such radio-frequency power source, the radio-frequency power is outputted only during the high level period of the pulse modulation control signals, and there is no output during the low level period of the signals. Accordingly, the resulting radio-frequency power has a pulsed form (see, for example, Patent Document 1).

In addition to the above-described ON/OFF control in which the output and non-output states of the radio-frequency power are switched, there is also known a control method in which the amplitude of the radio-frequency power is switched between two levels, i.e., a first level and a second level which is lower than the first level. In this case of two-level control, for example, the voltage supplied to the amplifier is subjected to the switching between the two levels, so that the power outputted from the amplifier has corresponding two levels switched to form a pulsed output.

PRIOR ART DOCUMENT Patent Document

Patent Document 1: JP-A-2013-135159

SUMMARY OF THE INVENTION Problems to be Solved by the Invention

It is difficult, however, to conduct high-speed switching of the voltage supplied to the amplifier, and hence to produce radio-frequency pulsed power corresponding to the desired high switching frequency (referred to as “pulse frequency” below) of the first and second levels. Likewise, due to the difficulty of high-speed changing of the voltage supplied to the amplifier, it is difficult to produce radio-frequency power having a desired waveform.

The present invention has been proposed under the above-noted circumstances, and an object of the invention is to provide a radio-frequency power source capable of outputting radio-frequency power of a desired fast-changing waveform.

Means to Solve the Problem

A radio-frequency power source includes; a radio-frequency generator that produces radio-frequency signals having a variable phase difference between them; a radio-frequency combiner that combines the radio-frequency signals outputted from the radio-frequency generator by a predetermined ratio depending on the phase difference, and that outputs to a load; an output controller that causes the radio-frequency generator to change the phase difference, thereby controlling radio-frequency power outputted the from radio-frequency combiner. The output controller performs control so that the phase difference changes so as to make the radio-frequency power outputted from the radio-frequency combiner into a desired waveform.

In a preferred embodiment of the invention, the output controller performs control so that the phase difference is switched between a first predetermined value and a second predetermined value.

In a preferred embodiment of the invention, the predetermined ratio is greater when the phase difference is equal to the first predetermined value than when the phase difference is equal to the second predetermined value.

In a preferred embodiment of the invention, the first predetermined value is equal to or greater than 0 [deg] and smaller than 90 [deg], and the second predetermined value is equal to or greater than 90 [deg] and equal to or smaller than 180 [deg].

In a preferred embodiment of the invention, the first predetermined value is equal to 0 [deg].

In a preferred embodiment of the invention, the second predetermined value is equal to 180 [deg].

In a preferred embodiment of the invention, the output controller performs feedback control with respect to the radio-frequency power by changing one of the first predetermined value or the second predetermined value.

In a preferred embodiment of the invention, the radio-frequency generator generates a first radio-frequency signal and a second radio-frequency signal, and the output controller switches a phase difference of the second radio-frequency signal relative to the first radio-frequency signal between the first predetermined value and the second predetermined value.

In a preferred embodiment of the invention, the output controller switches the phase difference among a first predetermined value, a second predetermined value and a third predetermined value.

In a preferred embodiment of the invention, the output controller changes the phase difference in accordance with a linear function.

In a preferred embodiment of the invention, wherein the output controller changes the phase difference in accordance with the following formula, where θ is the predetermined phase difference, and x(t) is a function corresponding to a desired waveform: θ=2·cos⁻¹(√x(t)).

In a preferred embodiment of the invention, the output controller switches the phase difference between a first predetermined value and a value of a predetermined function.

In a preferred embodiment of the invention, the output controller sets the phase difference to a predetermined phase difference at a time when power output to the load starts, where the power output becomes greater when the predetermined phase difference is set than when each of the first predetermined value and the second predetermined value is set.

In a preferred embodiment of the invention, the output controller does not set the predetermined ratio to zero.

In a preferred embodiment of the invention, the radio-frequency combiner is constituted by hybrid circuitry including a transmission transformer and a power-consuming resistor. When there is a phase difference between the plurality of radio-frequency signals, the resistor thermally consumes power corresponding to the phase difference, and the remaining power is outputted from the radio-frequency combiner.

Effect of the Invention

According to the present invention, by adjusting the phase difference, it is possible to change the waveform of radio-frequency power combined by and outputted from the radio-frequency combiner. Since the phase difference between the radio-frequency signals generated by the radio-frequency generator can be changed at high speed, it is possible to output radio-frequency power having a desired waveform changing at high speed.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the internal configuration of a radio-frequency power source of the present invention.

FIG. 2 shows an example of circuit of a DC-DC converter constituting a DC-DC converting circuit.

FIG. 3 shows an example of a DC-RF converting circuit.

FIG. 4 shows an example of hybrid circuitry constituting an RF combining circuit.

FIG. 5 shows the relationship between the power combining ratio of the RF combining circuit and the phase difference.

FIG. 6 shows an example of an RF combining circuit.

FIG. 7 shows the internal configuration of a radio-frequency signal generating circuit and illustrates a radio-frequency signal generating method.

FIG. 8 shows two radio-frequency signals outputted from the radio-frequency signal generating circuit.

FIG. 9 shows the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit.

FIG. 10 shows a block configuration example including three DC-RF converting circuits and two RF combining circuits.

FIG. 11 shows another block configuration example including three DC-RF converting circuits and two RF combining circuits.

FIG. 12 shows a block configuration example including four DC-RF converting circuits and three RF combining circuits.

FIG. 13 shows another block configuration example including four DC-RF converting circuits and three RF combining circuits.

FIG. 14 shows examples of circuits for constituting an RF combining circuit to combine the input of three or more powers.

FIG. 15 shows a configuration of a plasma processing system with an impedance matching device included.

FIG. 16 shows the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit.

MODE FOR CARRYING OUT THE INVENTION

Preferred embodiments of the present invention will be described below with reference to the accompanying drawings. In particular, as an example, a radio-frequency or RF power source applied to a plasma processing system is described.

FIG. 1 is a block diagram showing the internal configuration of a radio-frequency power source according to the present invention.

The radio-frequency power source 1 shown in FIG. 1 is configured to output pulsed radio-frequency power having a high level period in which the amplitude becomes a first level and a low level period in which the amplitude becomes a second level which is lower than the first level. The radio-frequency power source 1 includes two power amplifiers and a power combiner or combining circuit to combine the output powers from the two power amplifiers. The power combiner can change the combining ratio in accordance with the phase difference θ between the two input voltage signals, where the ratio is variable from a state in which all the inputted power is outputted to a state in which all the inputted power is thermally consumed, thereby resulting in zero output. The illustrated radio-frequency power source 1 is so configured that the phase difference θ of the two radio-frequency voltages v_(a), v_(b) inputted to the two power amplifiers is switched between two values (i.e., the first phase difference θ1 and the second phase difference θ2 (>θ1)), whereby desired pulsed radio-frequency power is outputted from the power combining circuit. Specifically, by setting the phase difference θ to the first phase difference θ1 for a certain period of time, the output from the power combining circuit becomes the first level of power (“high level period”). Further, by setting the phase difference θ to the second phase difference θ2 for the subsequent, certain period of time, the output from the power combining circuit becomes the second level of power (“low level period”). By repeating this, desired pulsed radio-frequency power is to be outputted.

The radio-frequency power source 1 includes an AC-DC converting circuit 2, a DC-DC converting circuit 3, a DC-RF converting circuit 4, an RF combining circuit 5, a filtering circuit 6, a power detecting circuit 10, a PWM signal generating circuit 7, a radio-frequency signal generating circuit 8, and a control circuit 9. A portion including the DC-RF converting circuit 4 and the RF combining circuit 5 constitutes radio-frequency generating circuitry U to output radio-frequency power to the load. The DC-RF converting circuit 4 includes two DC-RF converting circuits 4A, 4B of the same configuration. The power P_(A) outputted from the first DC-RF converting circuit 4A and the power P_(B) outputted from the second DC-RF converting circuit 4B are combined or synthesized by the RF combining circuit 5. The combined power is outputted to the load (a plasma processing apparatus, not shown) connected to an output terminal of the radio-frequency power source 1.

The AC-DC converting circuit 2 is a circuit block that generates an input voltage (DC voltage) V_(cc) to the DC-DC converting circuit 3 based on commercial power source. The AC-DC converting circuit 2 can be constituted by a well-known circuit. For example, the AC-DC converting circuit 2 may include a rectifying circuit in which four semiconductor rectifying elements are bridge-connected. The voltage inputted from the commercial power source is rectified by the rectifying circuit and smoothed by a smoothing circuit to produce DC voltage V_(cc).

The DC-DC converting circuit 3 converts the DC voltage V_(cc) inputted from the AC-DC converting circuit 2 into a DC voltage V_(dc) of a desired voltage value and outputs the converted DC voltage to the DC-RF converting circuit 4.

The DC-DC converting circuit 3 can be constituted by a well-known DC-DC converter shown in FIG. 2 in which a rectifying circuit is combined with an inverter. The circuit example of FIG. 2 includes an inverter 301 formed of a bridge circuit in which four semiconductor switch elements Q_(A) are bridge-connected, and a rectifying circuit 302 connected to the inverter 301 via a transformer T1. The rectifying circuit 302 includes four semiconductor rectifying elements DA bridge-connected and a smoothing capacitor C connected in parallel to a pair of output terminals. The two output terminals of the rectifying circuit 302 are respectively connected to two output ends a, a′ of the DC-DC converting circuit 3. As the semiconductor switch elements Q_(A), use is made of a bipolar transistor, a field effect transistor, an IGBT, or the like. As the semiconductor rectifying element D_(A), use is made of a diode.

The DC-DC converting circuit 3 switches the four semiconductor switching elements Q_(A) of the inverter 301 between ON state and OFF state, based on the PWM signal S_(PWM) inputted from the PWM signal generating circuit 7. A DC voltage V_(dc) corresponding to the duty ratio (“PWM duty ratio”) of the PWM signal SPWM is outputted from the DC-DC converting circuit 3. The larger the PWM duty ratio is, the larger the DC voltage V_(dc) becomes.

The DC-RF converting circuit 4 converts DC power input from the DC-DC converting circuit 3 into predetermined radio-frequency power. The output frequency of the radio-frequency power is chosen for plasma processing, such as 2.0 MHz or 13.56 MHz. In the DC-RF converting circuit 4, two DC-RF converting circuits 4A, 4B of the same configuration are provided.

The first and second DC-RF converting circuits 4A, 4B are configured by half-bridge type, class-D amplifiers shown in FIG. 3. The class-D amplifier of the figure has two identical type of semiconductor switching elements Q_(B) connected in series between a pair of power supply terminals b and b′. An output circuit 401 is connected between the connection point n of the two semiconductor switch elements Q_(B) and the output terminal c. The output circuit 401 is a filtering circuit including a direct-current cutting capacitor and an L-shaped circuit of a capacitor and a reactor. The transformer T2 constitutes a drive circuit for driving the two semiconductor switch elements Q_(B). In the transformer T2, radio-frequency voltage v is inputted to the primary winding, and radio-frequency voltage v′ having the same phase as the radio-frequency voltage v is outputted from a first secondary winding (the upper secondary winding in FIG. 3). Further, from the other secondary winding (the lower secondary winding in FIG. 3) is outputted radio-frequency voltage −v′ opposite in phase to the radio-frequency voltage v. The radio-frequency voltage v′ is inputted to the first semiconductor switch element Q_(B) (the upper semiconductor switch element Q_(B) in FIG. 3), and the radio-frequency voltage −v′ is inputted to the second semiconductor switch element Q_(B) (the lower semiconductor switch element Q_(B) in FIG. 3). The radio-frequency voltage v to be inputted to the primary winding of the transformer T2 is a sine wave voltage having an output frequency f for plasma processing, such as 2.0 MHz or 13.56 MHz.

The power supply terminals b and b′ of the first DC-RF converting circuit 4A are connected to the two terminals b and b′ of the second DC-RF converting circuit 4B, respectively. A DC voltage V_(dc) outputted from the output terminals a, a′ of the DC-DC converting circuit 3 is supplied between the power supply terminals b and b′. As the semiconductor switch element Q_(B), use may be made of an N-channel type MOSFET, but another type of transistor such as a bipolar transistor or the like can also be used. Alternatively, the paired semiconductor switch elements Q_(B) may be constituted as a complementary type, including one N-channel type and one P-channel type. In this case, use may be made of only a single secondary winding in the transformer T2, where the radio-frequency voltage v′ is inputted to the gates of the respective N channel type and P channel type MOSFETs.

In the first and second DC-RF converting circuits 4A and 4B, radio-frequency voltage v_(a) or v_(b) (subscripts “a” and “b” correspond to the first DC-RF converting circuit 4A and the second DC-RF converting circuit 4B, respectively) to be inputted to the primary winding of the corresponding transformer T2 is generated by the radio-frequency signal generating circuit 8. Specifically, the radio-frequency signal generating circuit 8 generates radio-frequency voltages represented by v_(a)=A·sin(ω·t+φ_(a)) and v_(b)=A·sin(ω·t+φ_(b)). Here, the angular frequency ω is equal to 2πf, and herein, the angular frequency ω may be used instead of the output frequency f. The initial phase φ_(a) of the radio-frequency voltage v_(a) is fixed to 0 [deg], while the initial phase φ_(b) of the radio-frequency voltage v_(b) is variable. The radio-frequency signal generating circuit 8 will change the initial phase φ_(b) (=θ) of the radio-frequency voltage v_(b) based on the phase difference θ=φ_(b)−φ_(a) inputted from the controlling circuit 9. The details of the manner to change the phase difference 9 will be described later. Alternatively, the initial phase φ_(a) may be varied while the initial phase φ_(b) may be fixed to 0 [deg], or both of the initial phases φ_(a), φ_(b) may be varied. For example, the initial phase φ_(a) may be varied in a range of 0 [deg] to −90 [deg], while the initial phase φ_(b) may be varied in a range of 0 [deg] to 90 [deg]. In this case, the phase difference θ=90 [deg] may correspond to a combination of φ_(a)=−45 [deg] and φ_(b)=45 [deg].

In the first DC-RF converting circuit 4A, when the radio-frequency voltage v_(a)=A·sin(ω·t) is inputted to the primary winding of the transformer T2, radio-frequency voltage v_(a)′ of the same phase, A′·sin(ω·t), is outputted from one of the secondary windings of the transformer T2. In addition, radio-frequency voltage −v_(a)′ of the reverse phase, −A′·sin(ω·t), is outputted from the other secondary winding of the transformer T2. The in-phase radio-frequency voltage v_(a)′ is inputted to one of the semiconductor switch elements Q_(B) (the upper semiconductor switch element Q_(B) in FIG. 3), and the reversed-phase radio-frequency voltage −v_(a)′ is inputted to the other semiconductor switch element Q_(B) (the lower side semiconductor switching element Q_(B) in FIG. 3). In the case where the two semiconductor switch elements Q_(B) are N-channel MOSFETs, one semiconductor switch element Q_(B) is turned on during a high level period of the radio-frequency voltage v_(a)′, and the other semiconductor switch element Q_(B) is turned on during a high level period of the radio-frequency voltage −v_(a)′. In this manner, the two semiconductor switch elements Q_(B) alternately and repeatedly are turned on or off for every half cycle of the radio-frequency voltage v_(a)′.

Since the two semiconductor switching elements Q_(B) are alternately and repeatedly turned on and off, as noted above, the voltage v_(n) at the connection point n becomes “V_(dc)” in the period of v_(a)′>0, and becomes the ground level in the period of v_(a)′≤0, and thus the changing of the voltage produces a rectangular waveform. The direct current component of the rectangular wave and the switching noise are removed by the output circuit 401, and a radio-frequency voltage v_(PA)=V·sin(ω·t), or amplified radio-frequency voltage v_(a), is outputted from the output terminals c and c′.

The second DC-RF converting circuit 4B is configured to operate in the same manner as the above-described first DC-RF converting circuit 4A, to output radio-frequency voltage v_(PB), or V·sin(ω·t+θ), corresponding to amplified radio-frequency voltage v_(b).

In the above embodiment, the first and second DC-RF converters 4A, 4B are provided by half-bridge-type amplifiers, but they may also be full-bridge-type or push-pull-type amplifiers. Further, the present disclosure is not limited to a switching amplifier, and use may be made of a linear amplifier.

The RF combining circuit 5 combines two radio-frequency powers P_(A), P_(B) outputted from the DC-RF converting circuit 4. The RF combining circuit 5 is constituted by, for example, a hybrid circuit including a transmission transformer T3 and a resistor R shown in FIG. 4. The hybrid circuit has one sum port N_(S) and two input ports N_(A), N_(B). When there is a phase difference between the AC voltage inputted to the input port N_(A) and the AC voltage inputted to the input port N_(B), part of the input power corresponding to the phase difference is thermally consumed by the resistor R and the remaining power is outputted.

As shown in FIG. 4, the radio-frequency voltage v_(PA) outputted from the first DC-RF converting circuit 4A is inputted to one input port N_(A), and the radio-frequency voltage v_(PB) outputted from the second DC-RF converting circuit 4B is inputted to the other input port N_(B). As a result, radio-frequency voltage v_(PX) is outputted from the sum port N_(S).

When the load connected to the sum port N_(S) has an impedance of R_(o)/2 (i.e., when the RF combining circuit 5 and the load are impedance-matched), the radio-frequency current i_(PX) and the radio-frequency voltage v_(PX) to be outputted from the sum port N_(S) are as follows, where the radio-frequency voltages v_(PA) and v_(PB) are V·sin(ω·t) and V·sin(ω·t+θ), respectively.

The voltage v_(R) across the resistor R is as follows. v _(R) =v _(PA) −v _(PB) =V·[sin(ω·t)−sin(ω·t+θ)]  (1) The currents i_(A), i_(B) inputted to the transmission transformer T3 from the input ports N_(A), N_(B) and the current i_(R) flowing through the resistor R are as follows.

$\begin{matrix} {i_{A} = {{V_{PA}/R_{o}} = {V \cdot {{\sin\left( {\omega \cdot t} \right)}/R_{o}}}}} & \; & (2) \\ {i_{B} = {{V_{PB}/R_{o}} = {V \cdot {{\sin\left( {{\omega \cdot t} + \theta} \right)}/R_{o}}}}} & \; & (3) \\ \begin{matrix} {i_{R} = {V_{R}/\left( {2 \cdot R_{o}} \right)}} \\ {= {V \cdot {\left\lbrack {{\sin\left( {\omega \cdot t} \right)} - {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack/\left( {2_{o} \cdot R_{o}} \right)}}} \end{matrix} & \; & (4) \end{matrix}$

Thus, the i_(LA), i_(LB) flowing through the primary winding and the secondary winding of the transmission transformer T3 are as follows. i _(LA) =i _(A) −i _(R) =V·[sin(ω·t)+sin(ω·t+θ)]/(2_(o) ·R _(o))  (5) i _(LB) =i _(B) +i _(R) =V·[sin(ω·t)+sin(ω·t+θ)]/(2_(o) ·R _(o))  (6) and, the radio-frequency current i_(PX) and radio-frequency voltage v_(PX) to be outputted from the sum port N_(S) are as follows.

$\begin{matrix} {i_{PX} = {{i_{LA} + i_{LB}} = {V \cdot {\left\lbrack {{\sin\left( {\omega \cdot t} \right)} + {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack/R_{o}}}}} & (7) \\ \begin{matrix} {V_{PX} = {i_{PX} \cdot \left( {R_{o}/2} \right)}} \\ {= {V \cdot {\left\lbrack {{\sin\left( {\omega \cdot t} \right)} + {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack/2}}} \\ {= {V \cdot {\left\lbrack {{\sin\left\{ {\left( {{\omega \cdot t} + {\theta/2}} \right) - {\theta/2}} \right\}} + {\sin\left\{ {\left( {{\omega \cdot t} + {\theta/2}} \right) + {\theta/2}} \right\}}} \right\rbrack/2}}} \\ {= {V \cdot \left\lbrack {{{\sin\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\cos\left( {\theta/2} \right)}} - {{\cos\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\sin\left( {\theta/2} \right)}} +} \right.}} \\ {\left. {{{\sin\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\cos\left( {\theta/2} \right)}} + {{\cos\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\sin\left( {\theta/2} \right)}}} \right\rbrack/2} \\ {= {V \cdot {\cos\left( {\theta/2} \right)} \cdot {\sin\left( {{\omega \cdot t} + {\theta/2}} \right)}}} \end{matrix} & (8) \end{matrix}$

The power P_(X) outputted from the output port N_(S) and the power P_(R) consumed by the resistor R are as follows.

$\begin{matrix} \begin{matrix} {{PX} = {{V_{PX}^{2}/\left( {R_{o}/2} \right)} = {2 \cdot {V_{PX}^{2}/R_{o}}}}} \\ {{= {V^{2} \cdot {\left\lbrack {{\sin\left( {\omega \cdot t} \right)} + {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)}}}} \\ {= {2 \cdot \left\lbrack {V \cdot {\cos\left( {\theta/2} \right)}} \right\rbrack^{2} \cdot {{\sin^{2}\left( {{\omega \cdot t} + {\theta/2}} \right)}/R_{o}}}} \end{matrix} & (9) \\ \begin{matrix} {P_{R} = {V_{R}^{2} \cdot \left( {2_{o}/R_{o}} \right)}} \\ {= {{V^{2} \cdot \left\lbrack {{\sin\left( {\omega \cdot t} \right)} - {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack^{2}}\left( {2_{o} \cdot R_{o}} \right)}} \\ {= {V^{2} \cdot {\left\lbrack {{\sin\left\{ {\left( {{\omega \cdot t} + {\theta/2}} \right) - {\theta/2}} \right\}} - {\sin\left\{ {\left( {{\omega \cdot t} + {\theta/2}} \right) + {\theta/2}} \right\}}} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)}}} \\ {= {V^{2} \cdot \left\lbrack {{{\sin\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\cos\left( {\theta/2} \right)}} - {{\cos\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\sin\left( {\theta/2} \right)}} -} \right.}} \\ {\left. {{{\sin\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\cos\left( {\theta/2} \right)}} - {{\cos\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\sin\left( {\theta/2} \right)}}} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)} \\ {= {V^{2} \cdot {\left\lbrack {{- 2}{{\cos\left( {{\omega \cdot t} + {\theta/2}} \right)} \cdot {\sin\left( {\theta/2} \right)}}} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)}}} \\ {= {2 \cdot \left\lbrack {V \cdot {\sin\left( {\theta/2} \right)}} \right\rbrack^{2} \cdot {{\cos^{2}\left( {{\omega \cdot t} + {\theta/2}} \right)}/R_{o}}}} \end{matrix} & (10) \end{matrix}$

The powers P_(A), P_(B) inputted from the input ports N_(A), N_(B) are P_(A)=V² sin²(ω·t)/R_(o) and P_(B)=V² sin²(ω·t+θ)/R_(o). Thus, the power P_(in) inputted to the RF combining circuit 5 is P _(in) =P _(A) +P _(B) =V ²·[sin²(ω·t)+sin²(ω·t+θ)]/R _(o) On the other hand, the total power P_(sum) obtained from the addition of the power P_(X) outputted from the RF combining circuit 5 and the power P_(R) thermally consumed by the resistor R is

$\begin{matrix} {P_{sum} = {P_{X} \times P_{R}}} \\ {= {{V^{2} \cdot {\left\lbrack {{\sin\left( {\omega \cdot t} \right)} + {\sin\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)}} + {V^{2} \cdot \left\lbrack {{\sin\left( {\omega \cdot t} \right)} -} \right.}}} \\ {\left. {\sin\left( {{\omega \cdot t} + \theta} \right)} \right\rbrack^{2}/\left( {2_{o} \cdot R_{o}} \right)} \\ {= {V^{2} \cdot \left\lbrack {{2{\sin^{2}\left( {\omega \cdot t} \right)}} + {2\;{{{s{in}}^{2}\left( {{\omega \cdot t} + \theta} \right)}/\left( {2_{o} \cdot R_{o}} \right)}}} \right.}} \\ {{= {V^{2} \cdot {\left\lbrack {{\sin^{2}\left( {\omega \cdot t} \right)} + {\sin^{2}\left( {{\omega \cdot t} + \theta} \right)}} \right\rbrack/R_{o}}}}{{Hence},{P_{in} = {P_{sum}.}}}} \end{matrix}$

Accordingly, when θ=0 [deg], then P_(R)=0 and the input power P_(in) itself is outputted, as output power P_(X), from the RF combining circuit 5. When θ=180 [deg], then PX=0 and the output from the RF combining circuit 5 is zero. When 0 [deg]<θ<180 [deg], the resultant power obtained by combining the input powers P_(A) and P_(B) by a predetermined ratio η(θ) depending on the phase difference θ is outputted as the output power P_(X) from the RF combining circuit 5.

The above ratio η(θ) is equal to cos²(θ/2), as shown in equation (9), and its graph is depicted in FIG. 5, indicated by (a). The power combining ratio η(θ) is 100% when the phase difference θ is 0 [deg]. As the phase difference θ increases, cos²(θ/2) monotonously decreases, and when the phase difference θ is 180 [deg], it becomes 0%. In the present embodiment, the phase difference θ is switched between a first phase difference θ1 (e.g., 20 [deg]) and a second phase difference θ2 (e.g., 160 [deg]). By switching between the larger combining ratio η(θ1) and the smaller combining ratio η(θ2), the output power P_(X) will become pulsed radio-frequency power. The reason why the first phase difference θ1 is set to 20 [deg] and the second phase difference θ2 is set to 160 [deg] is as follows. The output power control is performed, as described below, by varying the first and second phase differences θ1, θ2. Thus, to allow desired variation, a certain range allowing for the variation is given to the first and second phase differences θ1, θ2. Without any limitation, it may be arranged that the value of the first phase difference θ1 is selected from a range of 0 [deg] to 90 [deg], while the value of the second phase difference θ2 is selected from a range of 90 [deg] to 180 [deg].

In the present embodiment, the first and second phase differences θ1, θ2 are set within a range of 0 [deg] to 180 [deg]. Alternatively, each phase difference may be set, for example, within a range of 180 [deg] to 360 [deg], or within a range of 0 [deg] to −180 [deg].

FIG. 5(a) corresponds to the case where the impedance of the load connected to the sum port N_(S) is R_(o)/2. When the given impedance is other than R_(o)/2, the changing of the phase difference θ in a range of 0 [deg] to 180 [deg] enables controlling the magnitude of the power P_(X) to be outputted from the RF combining circuit 5.

The configuration of the hybrid circuit used for the RF combining circuit 5 is not limited to that shown in FIG. 4. For example, use may be made of a hybrid circuit having a configuration as shown in FIG. 6 for the RF combining circuit 5. In the hybrid circuit of FIG. 6, each end of the primary winding of the transmission transformer T3 is connected to one end of the secondary winding via a capacitor C′, and the four terminals, that is, both ends of the primary winding and both ends of the secondary winding are in unbalanced state. To be used as the RF combining circuit 5, one terminal p1 of the primary winding serves as an output terminal of combined power. The other terminal p2 of the primary winding and one terminal p3 of the secondary winding serve as input terminals, and the other terminal p4 of the secondary winding is connected to a resistor R for thermal consumption.

By the circuit configuration of FIG. 4, the power consumption P_(R) of the resistor R is zero when the phase difference θ is 0 [deg]. In the circuit configuration of FIG. 6, on the other hand, the power consumption P_(R) of the resistor R is zero when the phase difference θ is 90 [deg], and as the phase difference θ deviates from 90 [deg], power P_(R) corresponding to the deviation is consumed at the resistance R. Specifically, by the circuit configuration of FIG. 6, the power combining ratio η(θ) advances by 90 [deg] relative to that of the circuit configuration of FIG. 4. Hence, as shown by FIG. 5(b), the characteristic denoted by cos²(θ/2+π/2)=sin²(θ/2) is obtained. In this case, the first phase difference θ1 and the second phase difference θ2 may be set within a range of −90 [deg] to 90 [deg]. Alternatively, they may be set within a range of 90 [deg] to 270 [deg], for example.

The RF combining circuit 5 may be substituted by other circuitry as long as the same function as that of the above-described hybrid circuit is performed. For instance, use may be made of a radio-frequency power combiner disclosed in JP-A-2008-28923 or an output combining circuit disclosed in JP-U-H04-48715.

The filtering circuit 6 is, for example, a low-pass filter (LPF) provided by a n type circuit with two capacitors and one reactor. The filtering circuit 6 removes harmonics of the radio-frequency voltage v_(PX) and radio-frequency current i_(PX) outputted from the RF combining circuit 5, while also outputting the resultant fundamental wave component to the load. The filtering circuit 6 is not limited to the above-noted n-type circuit made up of capacitors and a reactor as long as it serves as a low-pass filter (LPF).

The power detecting circuit 10 may detect, without limitation, forward wave power P_(f) outputted from the radio-frequency power source 1. The power detecting circuit 10 includes a directional coupler, from which the power detecting circuit 10 detects the forward wave voltage v_(f) and the reflected wave voltage v_(r) included in the radio-frequency voltage v_(out). Then, the power detecting circuit 10 converts the forward wave voltage v_(f) into forward wave power P_(f) and outputs it to the controlling circuit 9. Alternatively, the reflected wave voltage v_(r) may be converted into reflected wave power Pr and outputted to the controlling circuit 9.

The PWM signal generating circuit 7 generates PWM signals S_(PWM) for driving the DC-DC converting circuit 3, and outputs them to the DC-DC converting circuit 3. The PWM signal generating circuit 7 generates the PWM signals S_(PWM) according to a preset PWM duty ratio. When it is necessary to increase the DC voltage V_(dc) outputted from the DC-DC converting circuit 3, the duty ratio is set to be an appropriately large value. When it is necessary to reduce the DC voltage V_(dc) outputted from the DC-DC converting circuit 3, the duty ratio is set to be an appropriately small value. As described later, the PWM duty ratio is set based on the target output power P_(fs1) of the high level period of the pulse. To this end, for example, a table or a relational formula defining the relationships between the target output power P_(fs1) and the PWM duty ratio may be previously given. Then, the PWM duty ratio can be set based on the table or the relational formula. As long as the target output power P_(fs1) is not changed, the PWM duty ratio is constant, and the DC voltage V_(dc) outputted from the DC-DC converting circuit 3 is also constant.

The radio-frequency signal generating circuit 8 generates the radio-frequency voltage v_(a) and the radio-frequency voltage v_(b), where the radio-frequency voltage v_(a) controls the driving of the semiconductor switch elements Q_(B) in the first DC-RF converting circuit 4A, and the radio-frequency voltage v_(b) controls the driving of the semiconductor switch elements Q_(B) in the second DC-RF converting circuit 4B. The radio-frequency signal generating circuit 8 generates the radio-frequency voltages v_(a), v_(b) based on information inputted from the controlling circuit 9, such as amplitude A, output frequency f and phase difference θ, while also outputting the radio-frequency voltage v_(a) to the first DC-RF converting circuit 4A, and the radio-frequency voltage v_(b) to the second DC-RF converting circuit 4B.

As shown in FIG. 7, the radio-frequency signal generating circuit 8 includes a first radio-frequency generating circuit 8 a for generating a sinusoidal radio-frequency voltage v_(a) and a second radio-frequency generating circuit 8 b for generating a sinusoidal radio-frequency voltage v_(b), where the radio-frequency voltage v_(b) is caused to have a phase difference θ with respect to the radio-frequency voltage v_(a) in response to the phase difference θ inputted from the controlling circuit 9. The first radio-frequency generating circuit 8 a and the second radio-frequency generating circuit 8 b are each provided by a direct digital synthesizer.

The following information regarding the radio-frequency voltage v_(a), that is, the amplitude A, the output frequency f, and the initial phase φ_(a) (=0 [deg]) are inputted from the controlling circuit 9 to the first radio-frequency generating circuit 8 a. As noted above, the output frequency f is 2.0 MHz or 13.56 MHz, for example, chosen for plasma processing systems. The initial phase φ_(a) can be set to an arbitrary value, but in the present embodiment it is set to 0 [deg]. Similarly, the following information regarding the radio-frequency voltage v_(b), that is, the amplitude A, the output frequency f, and the initial phase φ_(b) are inputted to the second radio-frequency generating circuit 8 b. Since θ=φ_(b)−φ_(a) and φ_(a)=0 [deg], the phase value θ outputted from the controlling circuit 9 is inputted as the information of the initial phase φ_(b). When φ_(a)≠0 [deg], a value (θ+φ_(a)) obtained by adding the initial phase φ_(a) to the phase difference θ outputted from the controlling circuit 9 is inputted as information of the initial phase φ_(b). The amplitude A and the output frequency f inputted to the second radio-frequency generating circuit 8 b are the same as the amplitude A and the output frequency f inputted to the first radio-frequency generating circuit 8 a. When the amplitude A and the output frequency f are to be fixed, the information regarding these fixed values may be preset in the first and second radio-frequency generating circuits 8 a, 8 b.

The first radio-frequency generating circuit 8 a generates a radio-frequency voltage v_(a) (digital signal; see v_(a) in FIG. 8) represented by A·sin(2πf·t)=A·sin(ω·t) based on the information of the amplitude A, the output frequency f and the initial phase φ_(a). Similarly, the second radio-frequency generating circuit 8 b generates a radio-frequency voltage v_(b) (digital signal; see v_(b) in FIG. 8) represented by A·sin(2πf·t)=A·sin(ω·t) based on the information of the amplitude A, the output frequency f and the control command value (θ).

The controlling circuit 9 controls the forward wave power P_(f) outputted from the radio-frequency power source 1 and the phase difference θ between the two radio-frequency voltages v_(a) and v_(b) generated by the first and second radio-frequency generating circuits 8 a and 8 b. The controlling circuit 9 is configured by a microcomputer including a CPU (Central Processing Unit), a ROM (Read Only Memory), and a RAM (Random Access Memory). The CPU executes control programs stored in the ROM to control the forward wave power P_(f) and the phase difference θ between the two radio-frequency voltages v_(a), v_(b).

The controlling circuit 9 receives the input of the pulse frequency of the pulsed radio-frequency power and the input of the duty ratio (“pulse duty ratio”) between the first and second levels of the pulsed radio-frequency power. This input may be initiated by the user using an input device (not shown) or initiated automatically by a preset program. In an embodiment, the pulse frequency (for example, 10 kHz) is lower (i.e., longer in cycle) than that of the radio-frequency voltages v_(a), v_(b), and the pulse duty ratio is, for example, 50%. Based on the pulse frequency and the pulse duty ratio, the controlling circuit 9 generates an output control signal for specifying the pulse waveform of the pulsed radio-frequency power. Then, the controlling circuit 9 switches the phase difference θ so that the phase difference becomes a first phase difference θ1 during the high level period of the output control signal and a second phase difference θ2 during the low level period of the output control signal.

When the phase difference θ becomes the first phase difference θ1 during the high level period of the output control signal, the phase difference θ between the radio-frequency voltages v_(a), v_(b) outputted from the radio-frequency signal generating section 8 becomes the same phase difference θ1. Also, the phase difference θ between the radio-frequency voltage v_(PA) outputted from the first DC-RF converting circuit 4A and the radio-frequency voltage v_(PB) outputted from the second DC-RF converting circuit 4B becomes the same phase difference θ1. Then, the output power P_(X) combined in accordance with the first phase difference θ1 is outputted from the RF combining circuit 5. In the present embodiment, the first phase difference θ1 is 20 [deg]. Thus, the output power P_(X) during the high level period is about 95% of P_(in), which is the sum of the power P_(A) outputted from the first DC-RF converting circuit 4A and the power P_(B) outputted from the second DC-RF converting circuit 4B (approximately 5% of the power P_(in) is thermally consumed by the RF combining circuit 5).

Likewise, when the phase difference θ becomes the second phase difference θ2 during the low level period of the output control signal, the phase difference θ between the radio-frequency voltages v_(a), v_(b) outputted from the radio-frequency signal generating section 8 becomes the same phase difference θ2. Also, the phase difference θ between the radio-frequency voltage v_(PA) outputted from the first DC-RF converting circuit 4A and the radio-frequency voltage v_(PB) outputted from the second DC-RF converting circuit 4B becomes the same phase difference θ2. Then, the output power P_(X) combined in accordance with the second phase difference θ2 is outputted from the RF combining circuit 5. In the present embodiment, the second phase difference θ2 is 160 [deg]. Thus, the output power P_(X) during the low level period is about 5% of the power P_(in) (approximately 95% of the power P_(in) is thermally consumed by the RF combining circuit 5).

In the above manner, the output power P_(X) outputted from the RF combining circuit 5 corresponds to pulsed radio-frequency power having a high level period which occupies about 95% of the power P_(in) and a low level period which occupies about 5% of the power P_(in).

FIG. 9 shows the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit 5. The radio-frequency voltage v_(PX) becomes high-level with a large amplitude when the phase difference θ is the first phase difference θ1, and becomes low-level with a small amplitude when the phase difference θ is the second phase difference θ2. In this manner, the radio-frequency power P_(X) outputted from the RF combining circuit 5 takes a pulse form.

Further, the controlling circuit 9 performs predetermined feedback control so that the radio-frequency power (forward wave power P_(f)) outputted from the radio-frequency power source 1 to the load is adjusted to become a target power. As the target power, a target output power P_(fs1) is set for the high level period, and a target output power P_(fs2) is set for the low level period. The user may manually input the target output powers P_(fs1) and P_(fs2) by operating an input device (not shown). Alternatively, the target output powers P_(fs1) and P_(fs2) may be automatically inputted by a program provided in advance.

During the high level period of the output control signal, the controlling circuit 9 calculates the deviation ΔP1 (=P_(fs1)−P_(f)) between the detected value of the forward wave power P_(f) inputted from the power detecting circuit 10 and the target output power P_(fS1). Also, based on the deviation ΔP1, the controlling circuit 9 generates a control command value for making the deviation ΔP1 zero. Then, the controlling circuit 9 changes the first phase difference θ1 based on the control command value, thereby controlling the forward wave power P_(f). In this manner, feedback control is performed so that the forward wave power P_(f) becomes the target output power P_(fs1). Likewise, during the low level period of the output control signal, the controlling circuit 9 calculates the deviation ΔP2 (=P_(fs2)−P_(f)) between the detected value of the forward wave power P_(f) inputted from the power detecting circuit 10 and the target output power P_(fs2). Also, based on the deviation ΔP2, the controlling circuit 9 generates a control command value for making the deviation ΔP2 zero. Then, the controlling circuit 9 changes the second phase difference θ2 based on the control command value, thereby controlling the forward wave power P_(f). In this manner, feedback control is performed so that the forward wave power P_(f) becomes the target output power P_(fs2).

In an embodiment, the control of the forward wave power P_(f) may be performed by changing the DC voltage V_(dc) outputted from the DC-DC converting circuit 3, instead of changing the first and second phase differences θ1, θ2. Specifically, the control command value generated by the controlling circuit 9 is outputted to the PWM signal generating circuit 7, and the PWM signal generating circuit 7 generates, based on the received control command value and a carrier signal generated by the PWM signal generating circuit 7, a PWM signal S_(PWM) using a triangular wave comparison method. Further, the output power control may be performed by arranging that the controlling circuit 9 changes, based on the control command value, the amplitude A outputted to the radio-frequency signal generating circuit 8.

As described above, in the radio-frequency power source 1 of the present embodiment, there are provided two DC-RF converting circuits, that is, the first DC-RF converting circuit 4A and the second DC-RF converting circuit 4B, together with an RF combining circuit 5 for combining the radio-frequency powers P_(A) and P_(B) of the respective DC-RF converting circuits 4A and 4B. In addition, the phase difference θ between the radio-frequency voltages v_(a) and v_(b) inputted to the first and second DC-RF converting circuits 4A and 4B is switched between the first phase difference θ1 and the second phase difference θ2. As a result, the output power P_(X) outputted from the RF combining circuit 5 is about 95% of the power P_(in) for the first phase difference θ1, and is about 5% of the power P_(in) for the second phase difference θ2. In other words, pulsed radio-frequency power having a high level period and a low level period is outputted. Since switching of the phase difference θ can be performed at high speed, the outputted, pulsed radio-frequency power has a high pulse frequency regarding the switching between the first level and the second level.

Further, in the radio-frequency power source 1 of the present embodiment, it is possible to output the pulsed radio-frequency power while the DC voltage V_(dc) outputted from the DC-DC converting circuit 3 is kept constant (when the target output power P_(fs1) is constant). Thus, no overshoot or undershoot occurs which would otherwise occur due to the change in the DC voltage V_(dc).

In the above embodiment, the case where the forward wave power P_(f) is controlled to follow or converge to the control target is described as an example, though the present disclosure is not limited thereto. For example, the radio-frequency power (forward wave power P_(f)−reflected wave power Pr) supplied to the load that may be controlled to follow or converge to a control target.

In the above embodiment, the first and second DC-RF converting circuits 4A, 4B of the same configuration are used for the DC-RF converting circuit 4, and the output powers P_(A), P_(B) of the DC-RF converting circuits 4A, 4B are combined by the RF combining circuit 5. Alternatively, use may be made of three or more DC-RF converting circuits, and their output powers may be combined together.

FIGS. 10-11 show a circuit configuration of a radio-frequency generating circuitry U′ provided with a DC-RF converting circuit 4′, including three DC-RF converting circuits of the same configuration, and an RF combining circuit 5′. The DC-RF converting circuit 4′ includes a third DC-RF converting circuit 4C in addition to first and second DC-RF converting circuits 4A, 4B, where the third DC-RF converting circuit 4C has the same configuration as the other two circuits. The RF combining circuit 5′ is provided with first and second RF combining circuits 5A, 5B of the same structure as the RF combining circuit 5.

The circuit configuration shown in FIGS. 10-11 may correspond to that in which the third DC-RF converting circuit 4C and the second RF combining circuit 5B are added to the DC-RF converting circuit 4 and the RF combining circuit 5 shown in FIG. 1, and in which the output powers of the combining circuit 5A and third DC-RF converting circuit 4C are combined by the second RF combining circuit 5B.

For providing three DC-RF converting circuits of the same configuration, use may be made of two methods as follows. According to a first method, the output voltages v_(PA), v_(PB) of the first and second DC-RF converting circuits 4A, 4B in the DC-RF converting circuit 4′ are driven with the phase difference θ=0, and the output voltage v_(PC) of the third DC-RF converting circuit 4C is driven with the phase difference θ with respect to the output voltages v_(PA) and v_(PB). According to a second method, the output voltage v_(PB) of the second DC-RF converting circuit 4B is driven with the phase difference θ with respect to the output voltage v_(PA) of the first DC-RF converting circuit 4A, and the output voltage v_(PC) of the third RF converting circuit 4C is driven with a phase difference ψ with respect to the output voltage v_(PX) of the first RF combining circuit 5A.

FIG. 10 shows the circuit configuration of the DC-RF converting circuit 4′ and the RF combining circuit 5′ for the first method. FIG. 11 shows the circuit configuration of the DC-RF converting circuit 4′ and the RF combining circuit 5′ for the second method.

According to the first method illustrated in FIG. 10, it is possible to replace the portion composed of the first and second DC-RF converting circuits 4A-4B and the first RF combining circuit 5A with an equivalent DC-RF converting circuit. Thus, the radio-frequency generating circuitry U′ is substantially the same as the above-described radio-frequency generating circuitry U (FIG. 1). Specifically, the first RF combining circuit 5A combines the output power P_(A) of the first DC-RF converting circuit 4A and the output power P_(B) of the second DC-RF converting circuit 4B with no power loss. Further, the second RF combining circuit 5B adjusts the output power P_(Z) to the load in accordance with the phase difference θ.

It is supposed that the waveforms of the radio-frequency signals v₁, v₂, v₃ inputted to the first, second and third DC-RF converting circuits 4A, 4B, 4C are represented by v₁=A₁·sin(ω·t+φ₁), v₂=A₂·sin(Ω·t+φ₂), v₃=A₃·sin(ω·t+φ₃). In the first method of FIG. 10, for example, a radio-frequency signal v_(a)=A·sin(ωt) (A₁=A₂=A, φ₁=φ₂=0) is inputted to the first and second DC-RF converting circuits 4A and 4B.

It is assumed that the input port and the output port of the RF combining circuits 5A and 5B are matched. Then, the output voltages v_(PA), v_(PB) of the first and second DC-RF converting circuits 4A, 4B are represented by v_(PA)=v_(PB)=V·sin(ω·t). Thus, by equation (8), the output voltage v_(PX) of the first RF combining circuit 5A is represented by V_(PX)=V·sin(ω·t). Therefore, when a radio-frequency signal v_(b)=A·sin(ω·t+θ) (A₃=A, φ₃=θ) is inputted to the third DC-RF converting circuit 4C, and v_(PC)=V·sin(ω·t+θ) is outputted from the third DC-RF converting circuit 4C, then the following output voltage v_(PZ) is obtained from the second RF combining circuit 5B. V _(PZ) =V·cos(θ/2)·sin(ω·t+θ/2)

The output powers P_(A) and P_(B) of the first and second DC-RF converting circuits 4A and 4B are combined by the first RF combining circuit 5A, without being thermally consumed. Thus, the power P_(X) (=P_(A)+P_(B)) is outputted from the first RF combining circuit 5A. In the second RF combining circuit 5B, the output power P_(X) and the output power PC of the third DC-RF converting circuit 4C are combined as shown in equation (9), and the following power P_(Z) is outputted. P _(Z)=2·[V·cos(θ/2)]²·sin²(ω·t+θ/2)/R _(o)

Thus, in the first method of FIG. 10, switching the phase difference θ between the first phase difference θ1 and the second phase difference θ2 makes it possible to change the combining amount for the output power P_(X) (=P_(A)+P_(B)) of the first and second DC-RF converting circuits 4A-4B and the output power PC of the third DC-RF converting circuit 4C, thereby producing the pulsed radio-frequency power P_(Z).

By the second method of FIG. 11, both the first RF combining circuit 5A and the second RF combining circuit 5B adjust the output power P_(Z) to the load. Supposing that the first and second DC-RF converting circuits 4A, 4B receive the inputs of radio-frequency signals, respectively, denoted by v_(a)=A·sin(ω·t) where φ₁=0 and v_(b)=A·sin(ω·t+θ) where φ₂=θ, and that the first and second DC-RF converting circuits 4A, 4B output voltages, respectively, denoted by v_(PA)=V·sin(ω·t) and v_(PB)=V·sin(ω·t+θ). In this situation, in light of equation (8), the voltage v_(PX) outputted from the first RF combining circuit 5A is expressed as follows. v _(PX) =V·cos(θ/2)·sin(ω·t+θ/2)

In addition, if it is arranged that a radio-frequency signal denoted by v_(C)=A₃·sin(ω·t+φ₃) is inputted to the third DC-RF converting circuit 4C, where A₃=A·cos(θ/2) and φ₃=θ/2+ψ, implying that the amplitudes A₃ and φ₃ are adjusted depending on the phase difference θ, and also that voltage v_(PC) denoted by V·cos(θ/2)·sin(ω·t+θ/2ψ) is outputted from the third DC-RF converting circuit 4C, then the following voltage v_(PZ) and power P_(Z) are outputted from the second RF combining circuit 5B. v _(PZ) =V·cos(θ/2)·cos(ψ/2)·sin(ω·t+θ/2+ψ/2) P _(Z)=2·[V·cos(θ/2)·cos(ψ/2)]²·sin²(ω·t+θ/2+ψ/2)/R _(o)

As noted above, according to the second method illustrated in FIG. 11, it is possible to output the power P_(Z) as pulsed radio-frequency power in two different modes. First, while fixing the phase difference ψ, the phase difference θ is switched between the first phase difference θ1 and the second phase difference θ2. Second, as opposite to the first mode, while the phase difference θ is fixed, the phase difference ψ is switched between ψ1 and ψ2. More specifically, by switching the phase difference θ between the first phase difference θ1 and the second phase difference θ2, the combining amount between the output power P_(A) of the first DC-RF converting circuit 4A and the output power P_(B) of the second DC-RF converting circuit 4B can be switched, thereby producing pulsed radio-frequency power as the power P_(Z). Alternatively, by switching the phase difference ψ between ψ1 and ψ2, it is possible to switch the combining amount between the combined power Px of the output powers P_(A), P_(B) from the first and second DC-RF converting circuits 4A, 4B and the output power Pc from the third DC-RF converting circuit 4C, thereby producing pulsed radio-frequency power as the power P_(Z).

FIGS. 12-13 show the circuit configuration of the radio-frequency generating circuitry U″ that includes a DC-RF converting circuit 4″ with four DC-RF converting circuits of the same configuration and an RF combining circuit 5″. In the DC-RF converting circuit 4″, there are additions of third and fourth DC-RF converting circuits 4C, 4D having the same configuration as the first and second DC-RF converting circuits 4A, 4B. The RF combining circuit 5″ is provided with a first RF combining circuit 5A, a second RF combining circuit 5B and a third RF combining circuit 5C each having the same configuration as the RF combining circuit 5.

The first RF combining circuit 5A of the RF combining circuit 5″ combines the output power P_(A) from the first DC-RF converting circuit 4A of the DC-RF converting circuit 4″ and the output power P_(B) from the second DC-RF converting circuit 4B. The second RF combining circuit 5B combines the output power P_(C) from the third DC-RF converting circuit 4C of the DC-RF converting circuit 4″ and the output power P_(D) from the fourth DC-RF converting circuit 4D. The third RF combining circuit 5C of the RF combining circuit 5″ combines the output power P_(X) from the first RF combining circuit 5A and the output power P_(Y) from the second RF combining circuit 5B.

There may be two methods practicable for providing four DC-RF converting circuits of the same configuration. In the first method, a phase difference θ is provided between the output voltage v_(PA) of the first DC-RF converting circuit 4A and the output voltage v_(PB) of the second DC-RF converting circuit 4B, as well as between the output voltage v_(PC) of the third DC-RF converting circuit 4C and the output voltage v_(PD) of the fourth DC-RF converting circuit 4D. This first method corresponds to providing two pairs of DC-RF converting circuit 4 and RF combining circuit 5 shown in FIG. 1 and combining the two powers outputted from the respective pairs.

FIG. 12 shows the circuit configuration of the DC-RF converting circuit 4″ and RF combining circuit 5″ related to the first method. Generally, the radio-frequency signals v₁, v₂, v₃, v₄ inputted to the first through fourth DC-RF converting circuits 4A, 4B, 4C, 4D have waveforms denoted by v₁=A₁·sin(ω·t+φ₁), v₂=A₂·sin(ω·t+φ₂), v₃=A₃·sin(ω·t+φ₃), and v₄=A₄·sin(ω·t+φ₄). By the first method, v₁=v_(a)=A·sin(ω·t) where A₁=A, φ₁=0, v₂=v_(b)=A·sin(ω·t+θ) where A₂=A and φ₂=θ, v₃=v_(a)=A·sin(ω·t) where A₃=A, φ₃=0, and v₄=v_(b)=A·sin(ω·t+θ) where A₄=A, φ₄=θ.

By the circuit configuration shown in FIG. 12, the first RF combining circuit 5A combines, by the prescribed ratio depending on the phase difference θ, the output power P_(A) from the first DC-RF converting circuit 4A and the output power P_(B) from the second DC-RF converting circuit 4B. Similarly, the second RF combining circuit 5B combines, by a prescribed ratio depending on the phase difference θ, the output power PC from the third DC-RF converting circuit 4C and the output power PD from the fourth DC-RF converting circuit 4D.

Assuming that the input ports of the RF combining circuits 5A, 5B and 5C are matched, the output power P_(X) from the first RF combining circuit 5A and the output power P_(Y) from the second RF combining circuit 5B are denoted as follows by equation (9). P _(X) =P _(Y)=2·V ²·cos²(θ/2)·sin²(ω·t+θ/2)/R _(o) Further, in the third RF combining circuit 5C, the output powers P_(X), P_(Y) are not thermally consumed. Thus, the third RF combining circuit 5C outputs the following output power P_(Z) to the load. P _(Z) =P _(X) +P _(Y)=4·V ²·cos²(θ/2)·sin²(ω·t+θ/2)/R _(o)

In the first method of FIG. 12, the switching of the phase difference θ between the first phase difference θ1 and the second phase difference θ2 makes it possible to change the combining amount of the output power P_(A) from the first DC-RF converting circuit 4A and the output power P_(B) from the second DC-RF converting circuit 4B, thereby outputting pulsed radio-frequency power as the power P_(X), while also outputting pulsed radio-frequency power as the power P_(Y) by changing the combining amount of the output power Pc from the third DC-RF converting circuit 4C and the output power PD from the fourth DC-RF converting circuit 4D. Then, the power P_(X) and the power P_(Y) are combined by the third RF combining circuit 5C and outputted as pulsed radio-frequency power for the power P_(Z).

According to the second method, the output voltage v_(PA) from the first DC-RF converting circuit 4A and the output voltage v_(PB) from the second DC-RF converting circuit 4B are controlled with the same phase. Similarly, the output voltage v_(PC) from the third DC-RF converting circuit 4C and the output voltage v_(PD) from the fourth DC-RF converting circuit 4D are controlled with the same phase. Further, a phase difference θ is provided between the output voltage v_(PX) from the first RF combining circuit 5A and the output voltage v_(PY) from the second RF combining circuit 5B.

FIG. 13 shows the circuit configuration of the DC-RF converting circuit 4″ and the RF combining circuit 5″ used for implementing the second method. According to the illustrated circuit configuration of FIG. 13, the first RF combining circuit 5A combines the output power P_(A) from the first DC-RF converting circuit 4A and the output power P_(B) from the second DC-RF converting circuit 4B with no power loss, and the second RF combining circuit 5B combines the output power Pc from the third DC-RF converting circuit 4C and the output power PD from the fourth DC-RF converting circuit 4D with no power loss. Then, the third RF combining circuit 5C combines the output power P_(X) from the first RF combining circuit 5A and the output power P_(Y) from the second RF combining circuit 5B at a predetermined ratio depending on the phase difference θ.

Supposing that the radio-frequency signals v₁, v₂ inputted to the first and second DC-RF converting circuits 4A, 4B are denoted by v₁=v₂=v_(a)=A·sin(ω·t), where A₁=A₂=A and φ₁=φ₂=0, the output voltage v_(PX) from the first RF combining circuit 5A is given as follows in light of equation (8). V _(PX) =V·sin(ω·t) Further, supposing that the radio-frequency signals v₃, v₄ inputted to the third and fourth DC-RF converting circuits 4C, 4D are denoted by v₃=v₄=v_(b)=A·sin(ω·t+θ), where A₃=A₄=A and φ₃=φ₄=θ, the output voltage v_(PY) from the second RF combining circuit 5B is as follows in light of equation (8). V _(PY) =V·sin(ω·t+θ)

Thus, the third RF combining circuit 5C outputs the following voltage v_(PZ) in light of equation (8). V _(PZ) =V·cos(θ/2)·sin(ω·t+θ/2)] Also, in light of equation (9), the following power v_(PZ) is outputted to the load. P _(Z)=2·[V·cos(θ/2)]²·sin²(ω·t+θ/2)/R _(o)

Thus, in accordance with the second method of FIG. 13, it is possible, by switching the phase difference θ between the first phase difference θ1 and the second phase difference θ2, to switch the combining amount of the output power P_(X) (=P_(A)+P_(B)) from the first RF combining circuit 5A and the output power P_(Y) (=P_(C)+P_(D)) from the second RF combining circuit 5B, thereby producing pulsed radio-frequency power as the power P_(Z).

In the embodiment of FIG. 1, the initial phase φ_(a) of the output voltage v_(PA) from the first DC-RF converting circuit 4A is fixed, while the initial phase φ_(b) of the output voltage v_(PB) from the second DC-RF converting circuit 4B is changed, and thus the phase difference θ=φ_(b)−φ_(a) is changed. Alternatively, the phase difference θ=φ_(b)−φ_(a) may be changed with the initial phase φ_(b) fixed while the initial phase φ_(a) is changed. Further, the change of the phase difference θ=φ_(b)−φ_(a) may be implemented by changing both the initial phases φ_(a) and φ_(b).

The above description of the embodiment relates to a circuit configuration in which the RF combining circuit 5 combines two RF powers. Alternatively, the RF combining circuit 5 may be configured to combine three or more RF powers. As a circuit configured for three or more RF powers, use may be made of circuits shown in FIG. 14.

For example, use may be made of a circuit shown in FIG. 14(b) for combining three RF powers. It is now assumed that the voltages inputted to the input terminals 1, 2, 3 are expressed as follows: v_(a)=A·sin(ω·t+φ_(a)), v_(b)=B·sin(ω·t+φ_(b)) and v_(C)=C·sin(ω·t+φ_(c)), and that their effective values are denoted by V_(arms), V_(brms) and V_(crms). Power P_(a)=V_(arms) ²/R, P_(b)=V_(brms) ²/R, and P_(c)=V_(crms) ²/R are inputted to the power combining circuit. If not v_(a)=v_(b)=v_(c), then differential voltages v_(ab)=v_(a)−v_(b), v_(bc)=v_(b)−vc, v_(ca)=v_(c)−v_(a) are applied to the three resistors R, respectively. Supposing that the effective values of the differential voltages v_(ab), v_(bc), v_(ca) are V_(abrms), V_(bcrms) and V_(carms), power P_(ab)=V_(abrms) ²/R, P_(bc)=V_(bcrms) ²/R, and P_(ca)=V_(carms) ²/R are thermally consumed by the three resistors R, respectively.

Thus, by providing phase differences θ_(ab), θ_(bc) and θ_(ca) between the input voltages v_(a), v_(b) and v_(c), it is possible to thermally consume a part (P_(ab)+P_(bc)+P_(ca)) of the inputted power P_(in)=P_(a)+P_(b)+P_(c), and to output the remaining power, P_(in)−(P_(ab)+P_(bc)+P_(ca)), from the power combining circuit to the load. The same applies to a case where four or more RF powers are inputted.

In the above embodiment, the output control of the radio-frequency power source 1 is described by taking the plasma processing system as an example, where a plasma processing apparatus is connected as the load to the radio-frequency power source 1. Alternatively, as shown in FIG. 15, the present invention may also be applied to a case where an impedance matching device 12 is provided between the radio-frequency power source 1 and the plasma processing apparatus 11.

When the impedance matching device 12 is provided, impedance matching between the radio-frequency power source 1 and the plasma processing apparatus 11 is performed by the impedance matching device 12 even if the impedance (load impedance) of the plasma processing device 8 fluctuates. However, in the transient period in which the impedance matching process by the impedance matching device 12 is being performed, the impedance mismatch can occur. Thus, even in the plasma processing system including the impedance matching device 12, the output control method for the radio-frequency power source 1 of the present invention is effective.

The above embodiment includes a radio-frequency generating circuitry U for combining a plurality of radio-frequency powers, and by switching the phase difference θ, for example, between the first phase difference θ1 and the second phase difference θ2, pulsed radio-frequency power with a high level period and a low level period is outputted. It should be note here that that gist of the above described techniques is not limited to a radio-frequency power source for a plasma processing system.

In the above embodiment, the radio-frequency voltage V_(out) outputted to the load has a sinusoidal waveform. Alternatively, it may have a trapezoidal waveform or a rectangular waveform with a dead time.

In the above embodiment, the phase difference θ outputted to the radio-frequency signal generating circuit 8 from the controlling circuit 9 is switched between two values θ1 and θ2, thereby switching the amplitude of radio-frequency power between the first level and the second level, for outputting pulsed radio-frequency power. The present invention is not limited to this. For example, the amplitude of the radio-frequency power may be switched among three or more levels.

The waveform shown in FIG. 16(a) corresponds to the case where the amplitude of the radio-frequency voltage v_(PX) outputted from the RF combining circuit 5 is switched among three levels. By switching the phase difference θ outputted to the radio-frequency signal generating circuit 8 from the controlling circuit 9 among three levels such as a first phase difference θ1 (For example 20 [deg]), a second phase difference θ2 (for example 90 [deg]), and a third phase difference θ3 (for example 160 [deg]), the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit 5 changes among three levels, as shown in FIG. 16(a). Accordingly, the radio-frequency power P_(X) outputted from the RF combining circuit 5 is switched in amplitude among three levels.

Instead of switching the phase difference θ among predetermined fixed values, the phase difference θ may be a function of time t, varying with time.

For example, let the phase difference θ be a linear function θ=a·t+b (a, b are constant) depending on time t. In this case, the combining ratio η(θ) in the RF combining circuit 5 takes the form shown in FIG. 5. Thus, the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit 5 is sinusoidal, as shown in FIG. 16(b). Hence, the radio-frequency power P_(X) outputted from the RF combining circuit 5 changes in a sinusoidal manner.

For changing the radio-frequency power P_(X) so as to take a desired waveform, the phase difference θ may be changed so that the radio-frequency voltage v_(PX) has a desired waveform. Since the combining ratio η(θ)=cos²(θ/2), the phase difference θ with respect to the combining ratio η is expressed by the following equation (11). θ=2·cos⁻¹(√η)  (11)

For example, when the radio-frequency voltage v_(PX) is to take the waveform (triangular waveform) shown in FIG. 16(c), the phase difference θ is caused to vary with time t so that the combining ratio η corresponds to the waveform shown in FIG. 16(c). To this end, in equation (11), the combining ratio η is substituted by the function x(t) representing the waveform of FIG. 16(c). In this manner, a desired combining ratio η can be set. For example, as in the waveform of FIG. 16(d), a triangular waveform and a constant-level waveform may be combined. As in the waveform shown in FIG. 16(e), a sinusoidal waveform and a constant-level waveform may be combined.

FIG. 16(b)-16(e), the combining ratio η(θ) in the RF combining circuit 5 may become zero at a time and therefore the output may be zero. When such zero output is not desired, the calculation formula of the phase difference θ may be adjusted so that the phase difference θ does not become 180 [deg].

As an example of the waveform of the radio-frequency voltage v_(PX), the waveform shown in FIG. 9 may be modified to obtain a waveform (see FIG. 16(f)) in which overshooting is implemented at the time of plasma ignition. To this end, in repeating a set of two periods, i.e., the first period t1 in which the phase difference θ is set to the first phase difference θ1 (for example 20 [deg]) and the second period t2 in which the second phase difference θ2 (for example 160 [deg]), a third period t3 for overshoot may be provided before the first period t1 at the time of plasma ignition. In such third period t3, the phase difference θ may be given by the following equation (12), where T is the length of the third period t3. In this manner, at the time of plasma ignition (at the start of the third period t3: t=0), the combining ratio η becomes maximum with the phase difference θ being 0. During the third period t3, the phase difference θ increases, while the combining ratio η decreases with time. At the end of the third period t3 (t=T), the phase difference θ becomes θ1. Note that the phase difference θ may be set to “0” at any time during the third period t3. By setting the radio-frequency voltage v_(PX) to the waveform including overshooting shown in FIG. 16(f), the radio-frequency voltage v_(out) outputted to the load becomes high when the plasma is not ignited. Hence, the ignitability of plasma can be improved. θ=(θ1/T)·t  (12)

It should be noted that the waveforms shown in FIG. 16 and the above calculation formulas including equation (12) are presented as mere examples. By appropriately setting the phase difference θ, the waveform of the radio-frequency voltage v_(PX) outputted from the RF combining circuit 5 can have various waveforms, and the waveform of the radio-frequency power PX outputted from the RF combining circuit 5 can have a desired waveform.

The radio-frequency power source according to the present invention is not limited to the above-described embodiments. The specific configuration of each part of the radio-frequency power source of the invention may be varied in many ways.

LIST OF REFERENCE CHARACTERS

-   -   1: Radio-frequency power source     -   2: AC-DC converting circuit     -   3: DC-DC converting circuit     -   4, 4′, 4″: DC-RF converting circuit (radio-frequency generator)     -   4A: First DC-RF converting circuit (radio-frequency generator)     -   4B: Second DC-RF converting circuit (radio-frequency generator)     -   4C: Third DC-RF converting circuit (radio-frequency generator)     -   4D: Fourth DC-RF converting circuit (radio-frequency generator)     -   401: Low pass filter     -   5, 5′, 5″: RF combining circuit (radio-frequency combiner)     -   5A: First RF combining circuit (radio-frequency combiner)     -   5B: Second RF combining circuit (radio-frequency combiner)     -   5C: Third RF combining circuit (radio-frequency combiner)     -   6: Filtering circuit     -   7: PWM signal generating circuit     -   8: Radio-frequency signal generating circuit (radio-frequency         generator)     -   8 a: First radio-frequency generating circuit     -   8 b: Second radio-frequency generating circuit     -   9: Controlling circuit (output controller)     -   10: Power detecting circuit     -   11: Plasma processing apparatus     -   12: Impedance matching device     -   U, U′, U″: Radio-frequency generating circuitry 

The invention claimed is:
 1. A radio-frequency power source comprising: a radio-frequency signal generator that produces radio-frequency signals having a variable phase difference between them; at least two amplifiers configured to amplify the radio-frequency signals, respectively; a voltage supplier configured to supply DC voltage to the at least two amplifiers; a radio-frequency combiner that combines amplified radio-frequency signals outputted from the at least two amplifiers by a predetermined ratio depending on the phase difference, and that outputs to a load; an output controller that causes the radio-frequency signal generator to change the phase difference, thereby controlling radio-frequency power outputted from the radio-frequency combiner, wherein the output controller performs control so that the phase difference changes so as to make the radio-frequency power outputted from the radio-frequency combiner into a desired waveform, while causing the radio-frequency signal generator to continue producing the radio-frequency signals when the DC voltage supplied from the voltage supplier remains constant.
 2. The radio-frequency power source according to claim 1, wherein the output controller performs control so that the phase difference is switched between a first predetermined value and a second predetermined value.
 3. The radio-frequency power source according to claim 2, wherein the predetermined ratio is greater when the phase difference is equal to the first predetermined value than when the phase difference is equal to the second predetermined value.
 4. The radio-frequency power source according to claim 3, wherein the first predetermined value is equal to or greater than 0 (deg) and smaller than 90 (deg), and the second predetermined value is equal to or greater than 90 (deg) and equal to or smaller than 180 (deg).
 5. The radio-frequency power source according to claim 4, wherein the first predetermined value is equal to 0 (deg).
 6. The radio-frequency power source according to claim 4, wherein the second predetermined value is equal to 180 (deg).
 7. The radio-frequency power source according to claim 1, further comprising a power detector configured to detect power outputted to the load, wherein the output controller performs feedback control by changing the phase difference so that the power detected by the power detector equals a target power.
 8. The radio-frequency power source according to claim 2, wherein the radio-frequency signal generator generates a first radio-frequency signal and a second radio-frequency signal, and the output controller switches a phase difference of the second radio-frequency signal relative to the first radio-frequency signal between the first predetermined value and the second predetermined value.
 9. The radio-frequency power source according to claim 1, wherein the output controller switches the phase difference among a first predetermined value, a second predetermined value and a third predetermined value.
 10. The radio-frequency power source according to claim 1, wherein the output controller changes the phase difference in accordance with a linear function.
 11. The radio-frequency power source according to claim 1, wherein the output controller switches the phase difference between a first predetermined value and a value of a predetermined function.
 12. The radio-frequency power source according to claim 2, wherein the output controller sets the phase difference to a predetermined phase difference at a time when power output to the load starts, and wherein the power output becomes greater when the predetermined phase difference is set than when each of the first predetermined value and the second predetermined value is set.
 13. The radio-frequency power source according to claim 1, wherein the output controller does not set the predetermined ratio to zero.
 14. The radio-frequency power source according to claim 1, wherein the radio-frequency combiner is constituted by hybrid circuitry comprising a transmission transformer and a power-consuming resistor, and wherein when there is a phase difference between the plurality of radio-frequency signals, the resistor thermally consumes power corresponding to the phase difference, and remaining power is outputted from the radio-frequency combiner. 